Tag: LHCP

Innovation: Mitigating interference with a dual-polarized antenna array in a real environment

Double Take

A diversely polarized antenna array combines signal processing in the spatial and polarization domains for significant improvement in receiver robustness against interference. The C/N₀ of line-of-sight components is improved since the receiver can use the power present in the left-hand circularly polarized channels, and also interference mitigation improves.

By Matteo Sgammini, Stefano Caizzone, Achim Hornbostel and Michael Meurer

INNOVATION INSIGHTS with Richard Langley

POLARIZATION. We use the word in everyday speech to mean a division into two groups with sharply contrasting opinions or beliefs.

But the word has another use in physics and electrical engineering to describe a characteristic of electromagnetic waves. Electromagnetic waves, whether they be light waves or radio waves, have electric and magnetic fields vibrating perpendicularly to each other and to the direction of propagation. If the electric field (and, correspondingly, the magnetic field) vibrates in a specific non-changing plane, we say that the wave is linearly polarized.

In terrestrial radio communications, signals are typically transmitted as linearly polarized waves with the electrical field oscillating in the vertical plane or the horizontal plane. Receiving antennas are designed and oriented to preferentially respond to the particular polarization of the signals. Before the widespread use of cable and satellite distribution platforms, VHF and UHF TV signals were received using rooftop antennas consisting of multiple parallel metal rods (similar antennas are used now for terrestrial digital TV).

In North America, the rods were in the horizontal plane since the transmitted signals were horizontally polarized. In Europe, on the other hand, the rods were sometimes in the vertical direction since there, some TV signals were transmitted with vertical polarization.

If the plane of vibration of the electric and magnetic fields rotates uniformly as the signal propagates, we have the case of circular polarization. Since the sense of rotation can be clockwise or anti-clockwise, we have right-hand circularly polarized (RHCP) and left-hand circularly polarized (LHCP) signals following the direction of curl of the fingers of the right and left hands. Circular polarization is typically used for signals from satellites in low and medium Earth orbit, such as GNSS satellites, where the relative orientation of the transmitting and receiving antennas is not fixed. For maximum signal reception, the polarization of the receiving antenna should match the polarization of the signal. All GNSS satellites transmit RHCP signals and therefore most GNSS receiving antennas are designed for such signals.

However, a funny thing can happen to a satellite signal on the way to a receiving antenna. If the signal bounces off a nearby structure or the ground or the sea surface, its polarization is modified and it will become LHCP or a combination of the two polarizations. While this multipath phenomenon can be a pest, as discussed in last month’s column, it can be used to advantage in measuring sea-surface roughness, for example, by monitoring reflected GNSS signals from a low Earth orbiting satellite or an aircraft using a LHCP antenna.

But GNSS receiving antennas are not perfect—especially for direct line-of-sight low-elevation-angle signals. A primarily LHCP antenna can capture a significant portion of the energy in such a RHCP signal and could provide a strong response to a reflected signal when the line-of-sight signal is missing or very weak. So, there could be a benefit in having a dual-polarized antenna to improve positioning capability in marginal situations. Furthermore, jamming signals can be of arbitrary polarization and a dual-polarized antenna array with beamforming capability could better separate and mitigate such interference. In this month’s column, we examine the principles of operation of such an antenna array and how one performed in real-world jamming and non-jamming scenarios.

The rapid growth of the wireless telecommunication sector and, consequently, the high demand of spectrum assigned to the new services make the frequency spectrum very crowded and quite saturated. With the weak received signal power of GNSS signals, spurious harmonics from other systems can cause unintentional interference and, therefore, a serious problem to the reliable estimation of user position, velocity and time (PVT). Besides unintentional interference, more virulent intentionally radiated signals, called jammers, may knock out the GNSS receiver; this is especially the case when a jammer with high time-frequency dynamics (such as a chirp-like jammer) affects the GNSS signal spectrum.

Whether unintentional or intentional, interference represents a serious threat to GNSS in applications ranging from safety-of-life to critical sectors like law enforcement, transportation, communication and finance. In such critical applications, it is important that the GNSS receiver provides a minimum level of reliability and robustness, even at the cost of increased price and complexity.

To meet this need, some manufacturers and research institutions have been developing GNSS receivers equipped with anti-jamming capabilities.

In this article, we propose a novel approach to interference mitigation. We equipped a GNSS receiver with a diversely polarized antenna array to combine signal processing in the spatial and polarization domains in a novel way. By doing this, we demonstrated achievable improvement in interference mitigation. For this purpose, we extended an existing two-step blind adaptive beamforming algorithm to a new algorithm that includes the polarization domain. We evaluated the new algorithm through measurement data gathered during a campaign carried out at the Automotive Testing Center in Aldenhoven, near Jülich, Germany. We used different interference sources, including low-cost jammers, euphemistically called personal privacy devices (PPDs), in real-life situations such as in a moving car approaching a GNSS receiver.

The receiving antenna used in our work is a four-element rectangular dual-polarized (DP) array in a two-by-two configuration. Each element has two feeds available, one ideally receiving the right-hand circularly polarized (RHCP) field and the other the left-hand circularly polarized (LHCP) field of the polarized incoming signals. Due to antenna imperfections and coupling effects, part of the LHCP field impinging on the antenna will be received by the RHCP port and vice versa. Generally, the antenna axial ratio is fine-tuned at boresight so that the energy of a RHCP satellite signal impinging on the array at high-elevation angles will be mostly captured by the RHCP port, while the energy flowing through the LHCP channel can be ignored. This statement does not hold for satellite signals coming from lower elevation angles or in general for signals with polarization other than RHCP, this being generally the case for multipath and interference. In particular, the response of a planar antenna array for angles-of-arrival (AoAs) close to the horizon is almost linearly polarized. It follows that a significant portion of the RHCP energy in a signal is likely to be captured by the LHCP channels and can be used either to strengthen the line-of-sight (LOS) component, or to better separate and mitigate both multipath and interfering (jamming) signals.

Adding the polarization domain makes it possible to better discriminate spatially and temporally correlated signals. In some environments, such as urban canyons, the LOS signal might not be available or might be strongly attenuated. In this case, the reflected non-LOS signals can be used to perform positioning and would benefit from a DP antenna approach. As a matter of fact, the reflected signals will be no longer RHCP, thus the LHCP channel can be used to strengthen the echoes and improve positioning. Diversely polarized antenna arrays also have the advantage of increasing the total number of available degrees of freedom. The number of degrees of freedom of an antenna array corresponds to the number of nulls that can be placed in the direction of arrival of interfering signals. For a single-polarization (SP) array with M elements, M-1 nulls can be placed in the spatial domain. In the case of a DP array, 2M-2 nulls can be placed in the space and/or polarization domains. This is a key factor in counteracting high-power and highly-dynamic jammers, such as PPDs. Furthermore, the use of a diversely polarized array improves signal detection, as well as direction of arrival and polarization-estimation performance. This is particularly true for closely spaced signals with sufficiently separated polarizations. On the other hand, the introduction of the second polarization increases the computational complexity of signal processing, since the number of elements is doubled.

The results of our measurement campaign show a significant improvement in receiver robustness against interference when the DP approach is used compared to the general SP case.

SIGNAL MODEL

In this section, we will briefly describe the theory of signal and antenna polarization with a minimal number of equations. A more complete discussion is included in the paper on which this article is based (see Further Reading).

Polarization of a Plane Wave. A received electromagnetic signal is assumed to be narrow band, and the source of radiation is assumed to be located in the far field. The plane wave propagating in free-space has the property that the direction of propagation is orthogonal to the electric and magnetic field vectors. This allows the electric field e of a polarized wave to be completely described in terms of the two unit vectors, and , orthogonal to the direction of propagation

(1)

where ∈_x and ∈_y are the real-valued, non-negative, amplitudes of the components of the electric field, Φ_x and Φ_y are the phase components of the field, ω is the angular frequency of the carrier and k is the wave number.

Only the real part of Equation (1) is physically relevant, with the complex exponential containing information about the phase of the oscillating field.

Switching from the linear to the circular basis vector set:

(2)

where and are the unit vectors of the RHCP and LHCP components, respectively and omitting the explicit time and spatial dependence, we can write the normalized electric field as

(3)

The polarization state of an electromagnetic signal is fully described by ∈^R and ∈^L.

More generally, the electric field of any plane wave impinging at the antenna can be expressed in the form

(4)

Dual-Polarized Antenna Array. A circular DP antenna features two orthogonal circular polarization output ports, meaning each element ideally receives the voltage induced by the RHCP and LHCP field components separately on the two different antenna ports. Due to antenna imperfections and the coupling effect, part of the received RHCP field is received by the LHCP port and vice versa. These undesired voltages are responsible for the emergence of the cross-polar components.

In view of this, we characterize the antenna in terms of its response to circularly polarized plane waves and express the electric field using the Jones vector notation in the circular basis as

(5)

where is the complex total electric field received by the RHCP port, is the complex electric field induced at the RHCP port by a purely RHCP electromagnetic wave, indicated as a co-polar component, is the complex cross-polar component of the electric field obtained by exciting the antenna with a purely LHCP electromagnetic wave, φ is the azimuth angle and θ is the elevation angle of the impinging signal assuming the antenna to be at the origin of the spherical coordinate system. Similar statements apply for the total electric field received by the LHCP port, and for the co-polar ( ) and cross-polar () components.

If v^R and v^L are the complex voltages induced at the RHCP and LHCP antenna outputs by the signal in Equation (4), respectively, the antenna outputs are given by

(6)

where ψ = [θ, φ] is the vector parameter carrying the information about the direction of arrival (DoA) of the incident signal and τ is the time delay of the incident signal.

With an M-element array of DP sensors, we can v^R and v^L to represent the complex array responses of the DP antenna array:

(7)

where and define the steering vector of the DP antenna array given a signal incident at angle ψ and polarization defined by the Jones vector .

Problem Formulation. The analog signals collected by the antenna array are then passed through the receiver front end where they are amplified, filtered and shifted to baseband. The resulting complex baseband signal with bandwidth B that is received by an antenna array with M DP sensor elements at polarization port P is

(8)

where s^p(t) defines the superimposed satellite signal replicas with l = 1 identifying the LOS signal and l = 2, …, L the non-LOS (multipath) signals and z^p(t) denotes the superimposed radio frequency interference (RFI) signals with i ranging from 1 to I.

Additionally, we assume temporally and spatially uncorrelated complex white Gaussian noise n^p(t). can be expressed in terms of the steering vectors given the lth signal’s incident angle, the polarization vector and a complex scalar term involving the signal complex amplitude, Doppler frequency, carrier-phase offset and the particular pseudorandom noise sequence and associated

The baseband signals are then digitized at sampling frequency 1/T_s≥ 2B. The observations are collected at K periods of the pseudorandom sequence at N time instances and the polarizations of the satellite signals and interferers as well as their DoAs are assumed to be constant over each single observation. We finally combine the two outputs of the DP antenna to benefit from both polarizations with a resulting unified signal output X. This increases the number of available degrees of freedom; furthermore, it allows us to carry out filtering in the polarization domain. On the other hand, the overall system complexity is increased; in particular, the computational complexity of the matrix inversion needed for pre-whitening (to be discussed next) is increased by a factor of about 2³.

PRE-WHITENING AND EIGEN-BEAMFORMING

Interference mitigation and beamforming uses a two-step blind beamforming approach based on orthogonal projection. It is similar to an approach we developed for the single-polarization case, with the only difference here being the introduction of the orthogonal LHCP channel, which doubles the number of sensors. Doubling the number of sensors does not necessarily mean that the number of degrees of freedom is also doubled. It has been shown that when using diversely polarized antennas, to discriminate signals unambiguously it is required that the maximum number of signals D = L + I satisfies the relationship D ≥ 2M–2.

This means that one additional degree of freedom is required to discriminate in the polarization domain in comparison to the case of an antenna array of uniformly polarized sensors, where it is required that D ≥ M–1.

Pre-Whitening. We establish a sample spatial-polarimetric covariance matrix, where we assume that the satellite signals, the interfering signals and the noise are uncorrelated among each other. Furthermore, we ignore the influence of the signal replicas, because their power is usually much smaller than the power of the noise and interference. We then obtain the approximate pre-whitening matrix to be applied to X. The pre-whitening matrix is applied before signal despreading.

Eigen-Beamforming. In the next stage, the complex pre-whitened signal passes through the tracking loops for despreading and code and carrier wipe-off. We collect the post-correlation signal at K integration intervals to obtain the data matrix and the post-correlation spatial-polarimetric sample covariance matrix. The post-correlation eigen-beamformer is obtained following the same optimization problem that we solved for the single-polarization case. We apply the optimum weight vector, maximizing the ratio between the power of the desired signal and the power of the undesired signals plus noise, using the eigenvector with respect to the dominant non-zero eigenvalues of the post-correlation covariance matrix.

MEASUREMENT CAMPAIGN

The receiving antenna used in our work is a planar four-element rectangular DP array in a two-by-two configuration, similar to one we have used previously, apart from the additional hybrid couplers needed to provide the LHCP channel outputs. Each element has a double feed, one ideally receiving the RHCP field and the other the LHCP field of the polarized incoming signals, resulting in a total of eight output channels. The single antenna elements are designed for the reception of the GPS L1 and L5 and Galileo E1 and E5 bands, but in this work we focus only on the reception of GPS L1 signals.

The eight signals are passed through a front end, where they are amplified, filtered and down-converted to the intermediate frequency of 2.5 MHz. The analog signals are then digitized with a sampling rate of 8 megasamples per second. The resulting 8-bit digital data are collected and stored on a solid-state drive for data analysis in post-processing. Data analysis is then performed by using a GNSS software-based receiver.

Description of Test Scenarios. The DP system has been tested using measurement data to assess its dual capability of improving the quality of LOS signal reception and robustness against both unintentional RFI and jamming. As mentioned previously, the measurement campaign was conducted at the Aldenhoven Automotive Testing Center. The location provides seven tracks of different lengths, inclinations and shapes. The test track used for this measurement campaign was the so-called autobahn, providing two lanes in each direction of travel and a total length of 1,000 meters (see FIGURE 1).

FIGURE 1. Test track layout.

In this article, we report and analyze the results of three different test scenarios. In the first test, we collected GPS L1 data over 60 seconds in an interference-free environment. The aim of this baseline scenario was to verify if the additional LHCP channels improved signal reception in terms of carrier-to-noise-density ratio (C/N₀) and PVT errors.

The second test scenario involved a horn antenna mounted on a mast, transmitting a continuous wave (CW) interference signal in the GPS L1 band and steered in the direction of the receiving antenna, as shown in FIGURE 2. Both the horn antenna and the receiving antenna were kept static during the measurement interval.

FIGURE 2. CW interference scenario.

The objective of the third test scenario was to replicate a real-life situation involving jamming, similar to the so-called “Newark scenario,” where a GPS jammer in a truck driving past Newark Liberty International Airport caused ground-based and satellite-based augmentation systems receivers to malfunction. To carry out this test, we installed a type K-320 PPD jammer transmitting in the GPS L1 band (see FIGURE 3) in the 12-volt auxiliary power outlet (cigarette lighter receptacle) of a moving car approaching the receiver and driving by.

FIGURE 3. The K-320 in-car PPD jammer.

The car started its run at a distance of about 260 meters from the receiver. During the first 20 seconds, the car holds its position. After this time, it was driven in the direction of the receiver with a constant speed of 30 kilometers per hour, finishing its route on the other side of the autobahn track, as depicted in Figure 1.

Baseline Scenario. The benefits that come to light using a DP array are of a dual nature. First, the C/N₀ of LOS signals is improved since the receiver can make use of the power present on the LHCP channels due to polarization mismatch, in particular for satellites with low AoA, resulting in better receiver-computed PVT solutions. This effect appears evident if we analyze the behavior of C/N₀ values over time collected during the non-interference experimental test in the GPS L1 band.

With reference to the sky plot in FIGURE 4 indicating the positions of the satellites at the time of observation, we analyzed the subgroup composed of those satellites having an elevation AoA lower than 30°. There was a sensible improvement of C/N₀using both polarizations from the DP antenna compared to just using the RHCP output (see FIGURE 5(a)). On the contrary, satellites with an elevation AoA higher than 60° do not benefit from the DP antenna and experienced almost the same C/N₀ whether the LHCP channel was used or not, as can be seen in FIGURE 5(b).

FIGURE 4. Receiver sky plot for GPS L1 on October 22, 2015, at 13:10:00 UTC.

FIGURE 5. C/N₀ improvement using the dual-polarized antenna: (a) low-elevation-angle satellites (elevation angle <30°), (b) high-elevation-angle satellites (elevation angle >60°).

While the advantage of using the DP array is evident when observing the C/N₀ behavior, this achievement does not translate with the same clear evidence when assessing the 2-D horizontal position error. Nevertheless, an improvement of about 6 centimeters in terms of the standard deviation of the 2-D position solution error in the horizontal plane has been obtained using the DP antenna (see TABLE 1). It is reasonable to expect that in a scenario where the availability of satellites is not as high as in our test case, the use of low-elevation angle satellites becomes more important for the accuracy of the PVT solution. In this case, the use of a DP antenna could play a key role in improving positioning accuracy.

Table 1. Interference-free RMS positioning error, in meters, in the horizontal plane over 60 seconds. Note that the data for the single-element result was obtained using just one sensor element of the 2 × 2 array in the same test run from which the array DP and array SP results were obtained.

CW Interference Scenario. The use of a DP array provides the ability to filter signals in the polarization domain, and at the same time we benefit from the additional degrees of freedom available. Thus, interference mitigation becomes more effective than using a SP array, increasing the receiver robustness and enabling tracking and successful PVT solutions in a severe interference scenario. This outcome appears evident analyzing the results of our test, where the linearly polarized CW interference described in TABLE 2 impinged on the array.

Table 2. Direction and calculated interference-to-signal ratio (ISR) for 25 dBm transmit power CW interference signal.

We show the 2-D horizontal position errors from this test in FIGURE 6. The figure highlights the improvement in position accuracy when both RHCP and LHCP channels are jointly used, limiting the root mean square (RMS) error to 2.65 meters, while it increases to 3.88 meters when only the RHCP channels have been used.

FIGURE 6. Horizontal position errors over 60 seconds in the presence of CW interference.

The advantages of using the DP array as assessed above are well summarized in FIGURE 7. The figure shows the history of the C/N₀split into two clusters. The upper cluster is from measurements during the interference-free period while the lower cluster is from measurements during the period the receiver is affected by the interference. In the figure, the improvement in terms of C/N₀ is notable when using the DP array, in particular for low-elevation angle satellites, and for those satellites having a DoA close to the DoA of the interference. The latter case, when satellite signals and the interfering signal almost overlap in space, has been fully analyzed in a technical note (see Further Reading).

FIGURE 7. C/N₀ history for all tracked GPS L1 satellites placed in order of their elevation AoA collected over 120 seconds: (a) using the single-polarized antenna, (b) using the dual-polarized antenna.

PPD Jammer Scenario. The goal of this test was to compare the overall performance of the DP array to the SP array, as well as to the case when only a single-element antenna was used and with no pre-whitening applied. The K-320 PPD employed in this test scenario poses a serious threat to any commercial receiver in obtaining a valid PVT solution. The spectrogram of the K-320 is shown in FIGURE 8(a), which illustrates that the chirp signal sweeps very rapidly (with a sweep interval of about 40 microseconds) across a frequency range of 15 MHz centered at the L1 carrier frequency, as can be seen in the plot of power spectral density in FIGURE 8(b). The frequency range is much larger than the receiver bandwidth of about 8 MHz (dual-sided). This means that the RFI is seen as pulsed RFI by the receiver.

FIGURE 8. Chirp-like signal generated by the K-320 PPD jammer: (a) spectrogram, (b) spectral density.

An estimate of the jamming behavior during the test in terms of interference-to-signal ratio (ISR) is shown in FIGURE 9. The estimated ISR counts only for the portion of jamming power falling within the receiver bandwidth in baseband after down conversion; it is not an estimate of the ISR at the antenna array. The closer the jammer in the passing car is to the receiver, the higher the PPD power affecting it. The minimum distance between the jammer and the receiver is about 14 meters and is reached at 13:13:45 UTC as indicated in the figure.

FIGURE 9. Estimated interference-to-signal ratio (ISR) of the K-320 PPD jammer.

In FIGURE 10, we can observe the impact of the RFI when the car drives past the receiver by means of the number of tracked satellites, or rather by the number of valid pseudoranges available for PVT computation. When the jammer is close to the receiver, the DP antenna is always better than the SP one. When the RFI is at the minimum distance (about 14 meters) from the receiver, the SP antenna is no longer able to deliver a valid position, while the DP antenna still can.

FIGURE 10. Number of available pseudoranges.

The higher number of valid pseudoranges when using the DP antenna is translated into a better position accuracy. This result can be seen in TABLE 3, which lists the RMS horizontal position error computed during the time the estimated ISR is greater than 25 dB. In the computations, only valid PVT solutions and 2-D positioning errors below 20 meters have been considered.

Table 3. RMS positioning error, in meters, in the horizontal plane computed when ISR > 25 dB.

CONCLUSION

The results of the measurement campaign have shown a significant improvement in positioning accuracy and robustness against interference when the dual-polarization approach is used compared to the general single-polarization case. Position accuracy takes advantage of the better C/N₀ for those satellites with an AoA below 30°, which experienced up to 2 dB C/N₀ improvement. Although the benefit in PVT accuracy was not remarkable in our testing, this should become more notable in scenarios where a lower number of satellites are visible or when the LOS signals are obstructed, such as in urban environments. Receiver robustness takes advantage of the possibility of filtering in the polarization domain and the additional number of available degrees of freedom, enabling tracking and PVT solution availability in severe interference scenarios. In particular, a valid PVT solution was still available for an ISR of 53 dB using the dual-polarization array, while the single-polarization array was unable to deliver a valid position. While these improvements are noteworthy, they do come with added cost and complexity of the receiving system, since the number of channels to be processed is doubled.

ACKNOWLEDGMENTS

This article is based on the paper “Interference Mitigation Using a Dual-Polarized Antenna in a Real Environment,” presented at ION GNSS+ 2016, the 29th International Technical Meeting of the Satellite Division of The Institute of Navigation, held Sept. 12–16, 2016, in Portland, Oregon.

MATTEO SGAMMINI received an M.Eng. degree in electrical engineering in 2005 from the University of Perugia, Italy. He joined the Institute of Communications and Navigation of the German Aerospace Center (DLR), Wessling, Germany, in 2008. He is pursuing a Ph.D. in electrical engineering with research interests in interference mitigation techniques for GNSS.

STEFANO CAIZZONE received an M.Sc. degree in telecommunications engineering and a Ph.D. degree in geoinformation from the University of Rome “Tor Vergata,” Italy, in 2009 and 2015, respectively. Since 2010, he has been with the antenna group of DLR’s Institute of Communications and Navigation, where he is responsible for the development of innovative miniaturized antennas.

ACHIM HORNBOSTEL holds a diploma degree in electrical engineering and a Ph.D. degree from the University of Hannover, Germany. He joined DLR in 1989 and heads a working group on algorithms and user terminals at the Institute of Communications and Navigation.

MICHAEL MEURER received a diploma in electrical engineering and a Ph.D. degree from the University of Kaiserslautern, Germany. Since 2006, he has been with DLR’s Institute of Communications and Navigation, where he is the director of the Department of Navigation and of the Center of Excellence for Satellite Navigation. Since 2013, he has also been a professor of electrical engineering and director of the Institute of Navigation at Rheinisch-Westfälischen Technischen Hochschule (RWTH) Aachen.

FURTHER READING

• Authors’ Conference Paper
“Interference Mitigation using a Dual-Polarized Antenna in a Real Environment” by M. Sgammini, S. Caizzone, A. Iliopoulos, A. Hornbostel and M. Meurer in Proceedings of ION GNSS+ 2016, the 29th International Technical Meeting of the Satellite Division of The Institute of Navigation, Portland, Oregon, Sept. 12–16, 2016, pp. 275–285.

• Technical Report on Overlapping Signals
Interference Mitigation using a Dual-Polarized Antenna:A Deep analysis in Space Domain and Polarimetric Domain by M. Sgammini. Internal Technical Report, Deutsches Zentrum für Luft- und Raumfahrt e. V. (DLR; German Aerospace Center), Dec. 2016.

• Authors’ Earlier Work
“Experimental Results of Interferer Suppression with a Compact Antenna Array” by A. Hornbostel, N. Basta, M. Sgammini, L. Kurz, S.I. Butt and A. Dreher in Proceedings of ENC-GNSS 2014, the European Navigation Conference, Rotterdam, The Netherlands, April 14–17, 2014.

“Detection and Suppression of PPD-Jammers and Spoofers with a GNSS Multi-Antenna Receiver: Experimental Analysis” by A. Hornbostel, M. Cuntz, A. Konovaltsev, G.C. Kappen, C. Hättich, C.A. Mendes da Costa and M. Meurer in Proceedings of ENC 2013, the European Navigation Conference, Vienna, Austria, April 23–25, 2013.

“Blind Adaptive Beamformer Based on Orthogonal Projections for GNSS” by M. Sgammini, F. Antreich, L. Kurz, M. Meurer and T.G. Nollin in Proceedings of ION GNSS 2012, the 25th International Technical Meeting of the Satellite Division of The Institute of Navigation, Nashville, Tennessee, Sept. 17–21, 2012, pp. 926–935.

“Field Test: Jamming the DLR Adaptive Antenna Receiver” by M. Cuntz, A. Konovaltsev, M. Sgammini, C. Hattich, G. Kappen, M. Meurer, A. Hornbostel and A. Dreher in Proceedings of ION GNSS 2011, the 24th International Technical Meeting of the Satellite Division of The Institute of Navigation, Portland, Oregon, Sept. 19–23, 2011, pp. 384–392.

“Suppression of Multipath and Jamming Signals by Digital Beamforming for GPS/Galileo Applications” by Z. Fu, A. Hornbostel, J. Hammesfahr and A. Konovaltsev in GPS Solutions, Vol. 6, No. 4, March 2003, pp. 257–264, doi: 10.1007/s10291-002-0042-2.

• Other Works on Antenna Beamforming
“GNSS Pest Control: Correlator Beamforming for Low-Cost Multipath Mitigation” by S. Gunawardena, J. Raquet and M. Carroll in GPS World, Vol. 28, No. 1, Jan. 2017, pp. 54–63.

“Null-Steering Antennas: Assessing the Performance of Multi-Antenna Interference-Rejection Techniques” by J.T. Curran, M. Bavaro and J. Fortuny-Guasch in GPS World, Vol. 27, No. 2, Feb. 2016, pp. 62–68.

• Diversely Polarized Antenna Arrays
“Subspace Fitting with Diversely Polarized Antenna Arrays” by A.L. Swindlehurst and M. Viberg in IEEE Transactions on Antennas and Propagation, Vol. 41, No.12, Dec. 1993, pp.1687–1694, doi: 10.1109/8.273313.

“Direction Finding with an Array of Antennas Having Diverse Polarizations” in IEEE Transactions on Antennas and Propagation, Vol. 31, No.2, March 1983, pp. 231–236, doi: 10.1109/TAP.1983.1143038.

• Antenna Array Signal Processing
“Two Decades of Array Signal Processing Research: The Parametric Approach” by H. Krim and M. Viberg in IEEE Signal Processing Magazine, Vol. 13, No. 4, July 1996, pp. 67–94, doi: 10.1109/79.526899.

“Multilinear Array Manifold Interpolation” by R.O. Schmidt in IEEE Transactions on Signal Processing, Vol.40, No.4, April 1992, pp. 857–866, doi: 10.1109/78.127958.

• Basic Antenna Concepts
“GNSS Antennas: An Introduction to Bandwidth, Gain Pattern, Polarization, and All That” by G.J.K. Moernaut and D. Orban in GPS World, Vol. 20, No. 2, Feb. 2009, pp. 42–48.

• GNSS Jamming
“Personal Privacy Jammers: Locating Jersey PPDs Jamming GBAS Safety-of-Life Signals” by J.C. Grabowski in GPS World, Vol. 23 No. 4, April 2012, pp. 28–37.

“GNSS Jamming in the Name of Privacy: Potential Threat to GPS Aviation” by S. Pullen and G.X. Gao in Inside GNSS, Vol. 7, No. 2, March/April, 2012, pp. 34–43.

“Know Your Enemy: Signal Characteristics of Civil GPS Jammers” by R.H. Mitch, R.C. Dougherty, M.L. Psiaki, S.P. Powell, B.W. O’Hanlon, J.A. Bhatti, and T.E. Humphreys in GPS World, Vol. 23, No. 1, Jan. 2012, pp. 64–72.

February 10, 2017
Interference mitigated with CRP and dual-polarized antennas: Free webinar

Two new topic areas and presentations have been added to this Thursday’s free webinar on Signal Interference: Detection and Mitigation.

The speakers will explore anti-jamming protection with controlled radiation pattern antennas (CRPAs) and with dual-polarized antennas. The latter topic is also the cover story for the February issue, which demonstrated a significant improvement in positioning accuracy and robustness against interference with a dual-polarization approach: a gain in terms of C/N₀, particularly for low-elevation angle satellites and valuable in urban environments.

Headshot: Kirk Burnell

Kirk Burnell from NovAtel joins the Feb. 2 panel to present “How to deliver assured positioning, navigation and timing in GNSS-compromised environments.”

He will look at applications that stress the importance of high-reliability PNT. Compromised GNSS signals due to unintentional interference is of great concern, but intentional interference due to jamming is much more insidious. Anti-jamming protection via controlled reception pattern antenna (CRPA) technology is now available to a wide range of users. A brief explanation of the technology will be followed by a few use-cases where CRPAs have been deployed in a variety of applications.

Burnell, Core Cards Product Manager for NovAtel, has worked at the company since 2015. With an education in survey engineering, Kirk has been working with precision GNSS system designers and integrators in both support and product management capacities for more than 20 years.

Headshot: Matteo Sgammini

Matteo Sgammini of the German Aerospace Center (DLR) will talk about work with dual-polarized antennas: the principles of operation of such an antenna array and how one performed in real-world jamming and non-jamming scenarios. This ION GNSS+ 2016 presentation became the cover story for GPS World’s February issue.

Innovation editor Richard Langley writes in his introduction to the February column, “All GNSS satellites transmit RHCP [right-hand circularly polarized] signals and therefore most GNSS receiving antennas are designed for such signals. However, a funny thing can happen to a satellite signal on the way to a receiving antenna. If the signal bounces off a nearby structure or the ground or the sea surface, its polarization is modified and it will become LHCP [left-hand circularly polarized] or a combination of the two polarizations.

“A primarily LHCP antenna can capture a significant portion of the energy in such a RHCP signal and could provide a strong response to a reflected signal when the line-of-sight signal is missing or very weak. So, there could be a benefit in having a dual-polarized antenna to improve positioning capability in marginal situations. Furthermore, jamming signals can be of arbitrary polarization and a dual-polarized antenna array with beamforming capability could better separate and mitigate such interference.”

February cover story. Photo: GNSS

Researchers at the DLR equipped a GNSS receiver with a diversely polarized antenna array to combine signal processing in the spatial and in the polarization domain. Tests show a significant improvement in receiver robustness against interference compared with the general single-polarization case.

The carrier-to-noise-density ratios of the line-of-sight components are improved since the receiver can use the power present on the left-hand circularly polarized channels, particularly for satellites with low elevation. Interference mitigation improves due to the possibility of filtering in the polarization domain and the additional number of available degrees of freedom.

Sgammini received a Masters degree in electrical engineering from the University of Perugia, Italy and now works at the Institute of Communications and Navigation, DLR. He is currently pursuing a Ph.D. in electrical engineering with research interests in interference mitigation techniques for GNSS. His research activity includes adaptive filtering, array signal processing and estimation theory for GNSS.

Sign up for this Thursday’s free webinar here.

Webinar Summary:

As the number of GNSS signals being tracked increases, so does the potential for interference to dismiss the performance gains of using those additional signals.

To maximize performance and efficiency, prepared PNT users need their equipment to be able to detect when interference is present and mitigate it.

Developers, integrators and users need mitigation tools to protect and preserve GNSS measurement quality, maintaining high-quality multi-frequency multi-constellation positioning performance, even in challenging RF environments. This is essential particularly on the integration journey, especially during prototyping and when encountering unforeseen interference events in field testing, in order to produce fully successful integrated products.

The one-hour webinar also will include a follow-up Q&A session with the speakers. Burnell and Sgammini join Patrick Casiano of NovAtel and Rick Hamilton of CGSIC on the speaker panel. Casiano will present an Interference Toolkit that measures RF spectrum levels and allows the user to apply mitigation tools to protect and preserve GNSS measurement quality. Hamilton will explain the proliferation of jammers, aspects of illegal use, coordinated government response to interference events, and regulations to prohibit manufacture, import, export, sale and use of jammers.

January 30, 2017
Innovation: Null-steering antennas

Assessing the performance of multi-antenna interference-rejection techniques

Several factors affect the levels of signal rejection using antenna arrays. Our authors describe experiments to assess the bounds the factors impose on its signal rejection capability.

By James T. Curran, Michele Bavaro and Joaquim Fortuny-Guasch

INNOVATION INSIGHTS with Richard Langley

IT’S ALL PHYSICS. How things work, that is. Well, maybe a little chemistry too in some cases. But I might be a little biased in my opinion given that I’m an applied physicist by training. Radio? Satellite navigation? Yes, the principles of their operation are all governed by physics. Many physicists of my generation started out as radio tinkerers. I’ve recounted in this column before that I built my first radio (from a kit) when I was 14 (not counting the crystal radio that my father helped me to put together when I was 9). Built a few more during high school, got into radio astronomy as an undergraduate, and did a Ph.D. in the application of very long baseline (radio) interferometry to geodesy.

The great American physicist Richard Feynman was also a radio tinkerer in his youth. He recounts in one of his autobiographical books how he used to fix radios. And because he would approach the task of repairing each non-functioning set by first contemplating why it wasn’t working, he got the reputation of fixing radios by thinking!

One of Feynman’s special abilities was in explaining how things worked. In fact, he has been called “The Great Explainer.” He authored what is arguably the best physics textbooks ever produced: The Feynman Lectures on Physics. The three-volume set, developed from his Caltech lectures to undergraduates between 1961 and 1964, covers mechanics, radiation, electromagnetism, matter and quantum mechanics. Many students and practicing physicists have learned or re-learned aspects of physics from the famous “red books.” Many more will now thanks to Caltech, which recently put the Lectures on line for anyone to read (feynmanlectures.caltech.edu).

In this month’s column, we are going to learn about the development of a microprocessor-controlled multi-element GNSS antenna array for interference rejection. While there are many textbooks that describe how multi-element antennas work, Feynman explains their operation in his Lectures from first principles — from the principles of physics.

The phenomenon governing the behavior of antennas with multiple elements is called interference. If we combine two electromagnetic waves, they will interfere with each other with a result that depends on the phase difference of the waves. The waves might reinforce each other leading to a larger net amplitude, called constructive interference, or partially or fully null each other out, called destructive interference. When we apply this concept to the signals received by a pair of antennas making up an array, we find that the array has directionality and we can have a null in the reception pattern in the directions parallel to the antenna baseline and will be insensitive to signals arriving from those directions. And as Feynman describes in his Lectures, by adding more antennas to the array and “some cleverness in spacing and phasing our antennas,” we can have a fairly narrow pattern null in a chosen direction. In the case of a GNSS antenna array, that direction might be that of a jamming signal and so we can null out the jammer and maintain a positioning capability.

Several factors affect the levels of signal rejection using antenna arrays. In this article, our authors describe these factors and the experiments they conducted with their microprocessor-controlled array to assess the bounds the factors impose on its signal rejection capability.

Directional antennas offer a powerful means of achieving signal selectivity when various signal sources observed by a receiver are separated spatially. In the context of GNSS, which must accommodate a mobile receiver observing many moving transmitters, adaptive antennas — or controlled radiation pattern antennas—are an attractive option. The benefits of antenna arrays have been demonstrated both for signal rejection, such as interference and multipath mitigation or anti-spoofing; and for the purposes of gain enhancement, angle-of-arrival, or attitude estimation.

A number of different factors can influence the achievable levels of signal rejection using antenna arrays. These factors include: the gain and phase stability of the analog radio-frequency (RF) and intermediate-frequency (IF) stages, the linearity of the active analog stages, and the fidelity of the signal-combining stages. Seeking to identify the bound imposed by each of these limiting factors, we have carefully examined the signal rejection capability of an antenna array in our work. The study considers a circular antenna array, consisting of seven passive dual-polarized (right-hand circularly polarized [RHCP] and left-hand circularly polarized [LHCP]) L1-L2 elements. Although signal rejection can be performed both in the analog and in the digital domain, this article focuses only on the analog combination of signals at RF, using a bank of controllable phase shifters and attenuators. We conducted broadcast experiments in a large-diameter anechoic chamber, housing a rotatable central pillar upon which the array is mounted, and two broadcast antennas mounted on movable sleds.

The results presented here include a precise three-dimensional phase and gain calibration of the antenna array using a network analyzer to explore the properties of antenna elements when placed in close proximity on a common ground plane. Further results include an investigation of the nulling depth achievable by the array via the synchronous broadcast of two GNSS-like code-division multiple access (CDMA) signals from different broadcast antennas. We then extrapolated these results to infer the relative degradation in nulling capability when the receiver’s estimate of the amplitude and phase of the signal to be rejected is poor. Finally, a comparison of analog and digital element combining is explored, with emphasis on the rejection of strong jamming signals.

This experiment sought to illustrate and quantify the unique benefits and limitations of each technique. In particular, we note that analog combining enjoys high linearity and can accommodate high interference power, but is typically restricted to the use of coarse phase and gain coefficients when combining elements. In contrast, digital combining can offer notably higher gain and phase resolution, but is limited by the dynamic range of the digitizer.

Antenna Characterization

The work reported in this article has focused on the use of a seven-element circular antenna array, consisting of dual-polarized (RHCP and LHCP), dual-frequency (L1 and L2) elements. The antenna elements are mounted on a single circular aluminum ground plane 2 millimeters thick and 50 centimeters in diameter, and placed in a hexagonal arrangement at a spacing of 12.5 centimeters, as depicted in FIGURE 1. Because the antennas are passive, and can be used both for transmission and for reception, characterization tests were performed in broadcast mode while the typical receive-mode operation of the array is performed using an in-line low-noise amplifier (LNA) after the antenna.

The experiments described here were conducted in an anechoic chamber, hemispherical in shape with a diameter of 20 meters, as depicted in FIGURE 2. The array was mounted on a surveyor’s tripod and placed at a known position on a rotatable pillar at the center of the chamber. The chamber contains two sleds, Sled A and B, which can be precisely positioned along an arc through the zenith at positions between ±115° either side of the vertical. These antennas include 1.0 to 6.0 GHz vertically and horizontally polarized standard-gain horn antennas.

FIGURE 2. Antenna array and digitizing front end in the anechoic chamber during broadcast tests.

Because the characteristics of the antenna array itself are central to the ultimate performance of beamforming or null-steering techniques, a thorough characterization of the gain and phase properties of each of the seven antenna elements was conducted.

To do so, a network analyzer was used to observe the gain and phase response of the antenna under test from a range of observation angles. The array was operated in transmit mode, broadcasting a signal sourced from Port A of the network analyzer, which was received by an antenna mounted on one of the movable sleds, and fed to Port B of the network analyzer.

The network analyzer was configured to broadcast a series of 201 equally spaced tones spanning 20 MHz centered at 1575.42 MHz at a power of -7 dBm from the antenna array.

A mechanical RF multiplexer was used to implement a time-division multiplexing of this broadcast measurement signal across each of the seven elements, such that the series of tones were transmitted once per antenna element. By performing the scan for each antenna element, for a range of positions of Sled A, and repeating this for different rotations of the central pillar, a precise frequency response could be calculated for a large set of points across the entire upper hemisphere of the antenna. The scan was computed on signals received by both the horizontal and vertical elements on Sled A, such that both the RHCP and LHCP response could be computed. The vertical cuts of this gain pattern were measured with resolution of 2°, while the horizontal cuts were measured with a resolution of 5°.

The average gain response, calculated across the 20-MHz band, for each of the seven elements is depicted in FIGURE 3. The elevation cut of the peripheral element is taken such that the -90° direction of the cut aligns with a radial line pointing away from the center of the array. The azimuth cuts are oriented such that the 0° direction aligns with a radial line extending from the center of element number 1 to the center of element number 2.

FIGURE 3. The measured gain pattern of the central element, number 1, (blue lines) and one of the peripheral elements, number 2, (red lines). The gain of the peripheral element is deflected inwards toward the center of the array because of the asymmetry of its positioning on the ground plane. (a) Elevation angle cut at an azimuth of 0°; (b) Azimuth cut at an elevation angle of 40°.

It is interesting to note that the gain pattern exhibited by each element is sensitive to its position on the ground plane and its position relative to other elements. Because of the rotational symmetry of the array, the gain patterns of all of the peripheral elements are similar, differing only in orientation, each one exhibiting a deflection of the maximum gain towards the center of the array. The central element is circularly symmetric with a single lobe in the direction of the zenith, while gain of the peripheral elements is deflected inwards, having lower gain away from the center of the array and an increased gain for high elevation angles from the center of the array. The difference in gain pattern across elements is stark and should, perhaps, influence the choice of elements to be used when forming a beam or null in a given direction. One or other of the signals should be scaled to compensate for this gain difference.

Measuring Signal Rejection

Before exploring factors that influence signal rejection, this section details the figure of merit, which might quantify the achievable performance of the array. We examined the nulling performance of the system in terms of its rejection capability: assessed as the relative received power of the signal of interest, b(t), that is to be preserved, and an unwanted signal, a(t), which is to be rejected, before and after the nulling combination. If s_j(t) denotes some signal as received at antenna j, then the combination of signals received at antennas j and k can be denoted by:

(1)

where κ and ϕ, respectively, represent a unitless scaling gain and a phase rotation in radians applied in the combination. When intending to form a beam in the direction of the source of s(t), then this phase might be chosen to bring s_k(t) into alignment with s_j(t), and the gain may be determined as a function of the signal-to-noise ratio at each antenna, or simply set to unity. In contrast, when it is intended to reject s(t) then eⁱ^ϕmust be chosen to place s_k(t) in antiphase with s_j(t) and must be chosen to scale the amplitude of s_k(t) to be exactly equal to that of s_j(t).

In this case, we consider the problem of placing a null in the direction of signal a(t) while preserving signal b(t). If the relative received power of a(t) and b(t) at antenna j is taken as a reference, then the rejection of a(t) with respect to b(t), denoted R_a_,_b, can be assessed by examining the change in relative power after the null has been placed:

(2)

where denotes the expected value of x. Note also that this convention implies that a value of R_a_,_bgreater than unity corresponds to signal rejection.

Analog Null Steering at RF

This section explores some of the receiver-side factors that can limit nulling performance. The performance of an analog RF-combining circuit is examined, wherein the combining function was implemented using controllable analog attenuators and phase shifters.

The received signal from each of two antennas, j and k, was fed to a custom RF circuit board hosting a controllable phase shifter and attenuator chips. The output of two of these boards was then combined using a passive power combiner, filtered by an analog RF filter, limiting the band to the range 1530–1620 MHz, and finally fed to a power detector, which produced a signal voltage that was proportional to the total observed power. The experimental setup is depicted in FIGURE 4. The attenuators and phase shifters were controlled digitally via a microcontroller board, which also sampled the output of the power detector.

FIGURE 5. A simplified example of the steering constellation of an analog gain and phase shifter, having 3-bit phase and gain control and a gain step-size of ~1 dB.

The attenuators accept a 6-bit control, providing a dynamic range of 30 dB in steps of approximately 0.5 dB, while the phase shifters accept a 4-bit control traversing the unit circle in steps of 22.5°.

A simplified example of the finite resolution achievable using such a phase and gain shifter is shown by the steering constellation depicted in FIGURE 5, taking the case of 3-bit gain and phase control and assuming a gain step size of 1 dB. Note that the gain is displayed on a logarithmic scale. Each of the circular markers represents a possible gain and phase coefficient for a received signal, which would be used to steer one signal, a, to be approximately equal in amplitude and in anti-phase with the second signal, b.

FIGURE 4. A custom-built programmable analog phase shifter and attenuator pair used for the analog null-steering configuration.

The residual misalignment between the signals stems from the finite constellation of steering points and results in a reduced nulling performance, whereby a portion of the interference signal remains. The relative magnitude of the remaining interference signal is maximum when the true relative phase and amplitude of the signals a and b lies equidistant from the four nearest steering vectors. This is depicted in Figure 5, where the cross marker lies equidistant from the four vertices located at the corners of {0°,45°} and {7,8} dB. Note that as the gain is depicted on a logarithmic scale, the relative error is equal for points centered in any of the quadrants.

To investigate the performance of the system, we broadcast a continuous-wave interference toward the array, while the signal from one antenna was manipulated by all possible gain and phase combinations, keeping the signal from the second antenna at a fixed zero phase shift and –15 dB attenuation. For each of the 1,024 possible gain and phase combinations, the power detector was sampled and logged. A trace of the measured signal rejection as a function of the gain and phase is depicted in FIGURE 6, wherein a sharp peak is observable at approximately {–15 dB, 210°}, corresponding to the point at which the unwanted signal is most rejected — in this particular case, to a level of approximately 29 dB.

FIGURE 6. The measured interference rejection for a broadcast jamming scenario, where a brute-force search through all possible combinations of phase shift and attenuation was conducted. In this case, the maximum rejection happens to occur at an attenuation of 16.5 dB and a phase shift of 225°.

Estimating the Achievable Rejection Level. In this particular experiment, because all 1,024 possible gain and phase combinations were examined in a brute-force search, the signal rejection was not limited by inaccuracies in the estimation of the steering variables κ and ϕ. Rather, it was limited by how accurately the steering variables can be applied. A residual error exists between the phase and gain that would perfectly align and null the signal and the nearest values of phase and gain that the circuit can produce. This error is a function of the distribution of the true steering parameter and the resolution with which it is rendered. In this case, as the range and angle to the unwanted signal source is arbitrary and the distance between antenna elements is comparable to the carrier wavelength, then it is reasonable, perhaps, to assume that the residual error in the steering parameters is zero mean and uniform over the discrete control steps. To model this effect, similar to the previous section, the combining function, inclusive of these errors, can be expressed as:

(3)

where U denotes a uniform distribution, δϕ denotes the step size of the phase shifter control and δA denotes the attenuator step size. Note that as κ is in units of amplitude and δA represents the discrete steps in power gain, which corresponds to discrete steps of in amplitude, then the residual error will be distributed over a region extending in either direction. In this case, if a B-bit phase shifter is used, then:

. (4)

From this model, the minimum expected rejection level can be estimated as a function of the phase and attenuator resolution. Considering first the rejection expression given by Equation (2), we note that the variation of the power signal of interest, b(t), is a function only of the relative angles between each of a(t) and b(t) and the antenna array. When the signals are well separated, a gain of 3 dB is observed on b(t), and when a(t) and b(t) are located nearby or in exact opposite directions, then the rejection of a(t) will also reject b(t). As this power variation is a function of geometry and not of the particular nulling technique, for simplicity it is assumed that b(t) experiences no power variation. What remains is the relative power variation of a(t) with respect to and δϕ.

To find the minimum expected rejection level, we must examine the following metric:

(5)

(6)

where the two variables, eκ and eϕ, respectively represent the residual errors in amplitude and phase between the perfect steering vector, and that which can be attained by the combiner. Examining Equations (3) and (6), it is clear that the minimum rejection will be achieved when the residual phase error is equal to e_ϕ = 1/2δϕ and the amplitude mismatch is given by e_κ = . Substituting these values yields the minimum expected rejection, as given in Equation (7):

.(7)

Determination of the average expected rejection level requires the averaging of Equation (6) over the distributions of the two error variables, e_κand e_ϕ. As these errors are assumed to be uniform in this particular case, this reduces to the following:

(8)

which, after some manipulation, admits the closed form expression of Equation (9):

.
(9)

Inserting the specifications of the experimental setup used here, we find that the minimum rejection that can be expected is equal to approximately 14 dB with an average value equal to 18.8 dB. Further exploring this result, it is possible to predict the minimum performance that can be achieved given some arbitrary, but finite, resolution in gain and phase rotation. A portion of the surface defined by Equation (9) is presented in FIGURE 7. One useful application of this result is that it may be used by a designer to ensure that the resolution in gain and in phase are commensurate. This can be inferred by examining the gradient of the surface, noting that optimal choices of gain and phase step size will lie along the line of steepest gradient of this surface. A flattening of the surface in one dimension indicates that the performance is limited by the other dimension. For example, it can be seen that an increase in phase resolution beyond 6 bits yields no improvement in rejection when the gain step size is greater than 0.5 dB.

FIGURE 7. Minimum achievable rejection of analog nulling-combiner as a function of phase-shifter resolution (bits) and attenuator step size (dB).

Conclusion

Early results from this study suggest that the achievable signal rejection using a controlled radiation pattern GNSS antenna, under ideal conditions, is in excess of 70 dB, and is primarily limited by the accuracy with which the angle of incidence of the interference can be estimated. Accounting for typical estimation errors, the nominal rejection levels of the order of 20 to 40 dB can be expected. However, it is observed that other aspects limit the signal rejection performance. In a practical receiver, these factors stem from component selection for the signal-combining circuitry.

For analog combining schemes, this is the resolution of the controlled attenuators and phase shifters used. The results here attempt to characterize the relationship between the minimum expected performance and the component properties. Results suggest that the choice of analog combining components should be chosen such that the phase and gain resolution are commensurate and such that resolution in one parameter is not rendered useless by a lack of resolution in the other. These results may form useful guidelines when designing analog RF null-steering antennas.

Acknowledgments

This article is based, in part, on the paper “Analog and Digital Nulling Techniques for Multi-Element Antennas in GNSS Receivers” presented at ION GNSS+ 2015, the 28th International Technical Meeting of the Satellite Division of The Institute of Navigation held in Tampa, Fla., Sept. 14–18, 2015.

Manufacturers

The equipment used in our study included an Agilent, now Keysight Technologies E8361A PNA network analyzer, Antcom Corporation 2DG1215A-MNS-4 GPS L1/L2 antennas, an Arduino LLC (www.arduino.cc) Arduino Uno microcontroller, a MACOM MAPS-010143 4-bit digital phase shifter, a Skyworks Solutions SKY12347-362LF 6-bit digital attenuator and a Tallysman Wireless TW127 in-line amplifier.

Further Reading

• Authors’ Conference Paper

“Analog and Digital Nulling Techniques for Multi-Element Antennas in  GNSS receivers” by J.T. Curran, M. Bavaro and J. Fortuny in Proceedings of ION GNSS+ 2015, the 28th International Technical Meeting of the Satellite Division of The  Institute of Navigation, Tampa, Fla., Sept. 14–18, 2015, pp. 3249–3261.

• Adaptive GNSS Antennas for Interference Suppression

“Advances in the Theory and Implementation of GNSS Antenna Array Receivers” by P. Arribas, C. Closas, M. Fernández-Prades, M. Cuntz, M. Meurer and A. Konovaltsev, Chapter 9 in Microwave and Millimeter Wave Circuits and Systems:  Emerging Design, Technologies, and Applications, edited by A. Georgiadis, H. Rogier, L. Roselli and P. Arcioni and published by Wiley, 2012, pp. 227–273.

“Mitigation of Continuous and Pulsed Radio Interference with GNSS Antenna Arrays” by A. Konovaltsev, D.S. De Lorenzo, A. Hornbostel and P. Enge in Proceedings of ION GNSS 2008, the 21st International Technical Meeting of the Satellite Division of The Institute of Navigation, Savannah, Ga., Sept. 16–19, 2008, pp. 2786–2795.

“Navigation Accuracy and Interference Rejection for an  Adaptive GPS Antenna Array” by D.S. De Lorenzo, J. Rife, P. Enge and D.M. Akos in Proceedings of ION GNSS 2006, the 19th International Technical Meeting of the Satellite Division of The Institute of Navigation, Fort Worth, Texas, Sept. 26–29, 2006, pp. 763–773.

“A Novel Interference Suppression Scheme for Global Navigation Satellite Systems Using Antenna Array” by M.G. Amin and W. Sun in IEEE Journal on Selected Areas in Communications, Vol. 23, No. 5, May 2005, pp. 999–1012, doi: 10.1109/JSAC.2005.845404.

“Wideband Cancellation of Interference in a GPS Receive Array” by R.L. Fante and J. Vaccaro in IEEE Transactions on Aerospace and Electronic Systems, Vol. 36, No. 2, April 2000, pp. 549–564, doi: 10.1109/7.845241.

• GNSS Antennas

“GNSS Antennas: An Introduction to Bandwidth, Gain Pattern, Polarization, and All That” by G.J.K. Moernaut and D. Orban in GPS World, Vol. 20, No. 2, February 2009, pp. 42–48.

“A Primer on GPS Antennas” by R.B. Langley in GPS World, Vol. 9, No. 7, July 1998, pp. 50–54.

JAMES T. CURRAN received a B.E. in electrical and electronic engineering in 2006 and a Ph.D. in telecommunications in 2010 from the Department of Electrical Engineering, University College Cork, Ireland. He worked as a senior research engineer with the Position, Location and Navigation group at the University of Calgary between 2011 and 2013 and is currently a grant holder at the Joint Research Center (JRC) of the European Commission (EC), Ispra, Italy. His main research interests are signal processing, information theory, cryptography and software-defined radios (SDRs) for GNSS.

MICHELE BAVARO received his master’s degree in computer science in 2003 from the University of Pisa, Italy. Shortly afterwards, he started his work on SDR technologies applied to navigation. First in Italy, then in The Netherlands and in the United Kingdom, he worked on several projects directly involved with the design, manufacture, integration, and test of GNSS equipment and supporting customers in the development of their applications. Today he is appointed as a grant holder at the EC JRC.

JOAQUIM FORTUNY-GUASCH received the engineering degree in telecommunications from the Technical University of Catalonia, Barcelona, Spain, in 1988, and the Dr.- Ing. degree in electrical engineering from the Universität Karlsruhe, Germany, in 2001. Since 1993, he has been working for the EC JRC as a senior scientific officer. He is the head of the European Microwave Signature Laboratory and leads the JRC research group on GNSS and wireless communications systems.

February 11, 2016